1. Field of the Invention
The present invention relates to a distortion compensation apparatus, and more particularly a distortion compensation apparatus which obtains a differential signal between a reference signal, i.e. a transmission signal, and a feedback signal, calculates a distortion compensation coefficient to reduce the differential signal by use of an adaptive algorithm, updates a stored distortion compensation coefficient using the above-calculated distortion compensation coefficient, and performs distortion compensation to the transmission signal based on the distortion compensation coefficient obtained above. In particular, preferably, the present invention relates to a digital predistortion device updating the stored data in a lookup table (LUT) having distortion compensation coefficients.
2. Description of the Related Art
In recent years, high-efficient digital transmission has been adopted in the radio communication field. When multilevel phase modulation is adopted in the radio communication, a technique for reducing adjacent channel leak power becomes important, in which nonlinear distortion is restrained by linearizing the amplification characteristic of a power amplifier on the transmission side.
Also, to improve power efficiency even in case an amplifier having a degraded linearity is used, a technique for compensating nonlinear distortion for the degraded linearity is necessary.
FIG. 1 shows an exemplary block diagram of transmission equipment in the conventional radio equipment. A transmission signal generator 1 outputs a digital serial data sequence. Also, a serial-to-parallel (S/P) converter 2 converts the digital data sequence into two series, in-phase component (I-component) signals and quadrature component (Q-component) signals, by alternately distributing the digital data sequence on a bit-by-bit basis.
A digital-to-analog (D/A) converter 3 converts the respective I-signals and Q-signals into analog baseband signals, and inputs the signals into a quadrature modulator 4. This quadrature modulator 4 performs orthogonal transformation and outputs signals by multiplying the input I-signals and Q-signals (transmission baseband signals) by a reference carrier wave 8 and a carrier wave phase-shifted therefrom by 90°, respectively, and adding the multiplied results.
A frequency converter 5 mixes the quadrature modulation signals with local oscillation signals, and converts the mixed signals into radio frequency. A transmission power amplifier 6 performs power amplification of the radio frequency signals output from frequency converter 5, and radiates the signal to the air from an antenna 7.
Here, in the mobile communication using W-CDMA, etc., transmission equipment power is substantially large, becoming as much as 10 mW to several tens of mW, and transmission power amplifier 6 has a nonlinear input/output characteristic having a distortion function f(p), as shown by the dotted line in FIG. 2. This non-linearity causes a non-linear distortion. As shown by the solid line (b) in FIG. 3, the frequency spectrum in the vicinity of a transmission frequency f0 comes to have a raised sidelobe from the characteristic shown by the broken line (a). This leaks to adjacent channels and produces adjacent interference. Namely, due to the nonlinear distortion shown in FIG. 2, leak power of the transmission wave to the adjacent frequency channels becomes large, as shown in FIG. 3.
An ACPR (adjacent channel power ratio) is used to indicate the magnitude of leak power. ACPR is a ratio of leak power to adjacent channels to the power in the channel of interest, in other words, a ratio of the spectrum area in the adjacent channels sandwiched between the lines B and B′ in FIG. 3 to the spectrum area between the lines A and A′. Such leak power affects other channels as noise, and degrades communication quality of the channels concerned. Therefore, a strict regulation has been established to the issue of leak power.
The leak power is substantially small in a linear region of, for example, a power amplifier (refer to a linear region I in FIG. 2), but is large in a nonlinear region II. Accordingly, to obtain a high-output transmission power amplifier, the linear region I has to be widened. However, for this purpose, it becomes necessary to provide an amplifier having a larger capacity than is actually needed, which causes disadvantage in apparatus cost and size. As a measure to solve this problem, a distortion compensation function to compensate for transmission power distortion is added to radio equipment.
FIG. 4 shows the block diagram of transmission equipment having a digital nonlinear distortion compensation function by use of a DSP (digital signal processor). A digital data group (transmission signals) transmitted from transmission signal generator 1 is converted into two series, I-signals and Q-signals, in S/P converter 2, and then the two series of signals are input to a distortion compensator 9.
As shown in the lower part of FIG. 4 in enlargement, distortion compensator 9 includes a distortion compensation coefficient storage 90 for storing a distortion compensation coefficient h(pi) corresponding to the power level pi(i=0−1023) of a transmission signal x(t); a predistortion portion 91 for performing a distortion compensation process (predistortion) onto the transmission signal, using the distortion compensation coefficient h(pi) corresponding to the transmission signal power level; and a distortion compensation coefficient calculator 92 for comparing the transmission signal x(t) with a demodulation signal (a feedback signal) y(t) demodulated in the quadrature detector which will be described later, and calculates and updates the distortion compensation coefficient h(pi) so that the difference between the transmission signal and the demodulation signal becomes zero.
The signal to which distortion process is performed in distortion compensator 9 is input into D/A converter 3. D/A converter 3 converts the input I-signal and Q-signal into analog baseband signals, and inputs the converted signals into quadrature modulator 4. Quadrature modulator 4 performs quadrature modulation by multiplying the input I-signal and Q-signal by a reference carrier wave 8 and a carrier wave being phase-shifted from carrier wave 8 by 90°. Quadrature modulator 4 then adds and outputs the multiplied result.
A frequency converter 5 mixes the quadrature modulation signal with a local oscillation signal, and performs frequency conversion. A transmission power amplifier 6 performs power amplification of the radio frequency signal output from frequency converter 5, and radiates the signal to the air by an antenna 7.
A portion of the transmission signal is input to a frequency converter 11 via a directional coupler 10, and input into a quadrature detector 12 after being converted by the above frequency converter 11. Quadrature detector 12 performs quadrature detection by multiplying the input signal by a reference carrier wave, and by a signal which is phase shifted by 90° from the reference signal, respectively. Thus, the baseband I-signal and Q-signal on the transmission side are reproduced, which are then input into an analog-to-digital (A/D) converter 13.
A/D converter 13 converts the input I-signal and Q-signal into digital signals, and inputs into distortion compensator 9. Through the adaptive signal processing, using an LMS (least-mean-square) algorithm, in distortion compensation coefficient calculator 92 of distortion compensator 9, the pre-compensated transmission signal is compared with the feedback signal being demodulated in quadrature detector 12. Then distortion compensator 9 calculates the distortion compensation coefficient h(p1) so as to make the above difference zero. Then, distortion compensator 9 updates the above-obtained coefficient which has been stored in distortion compensation coefficient storage 90. Through the repetition of calculations above, nonlinear distortion in transmission power amplifier 6is restrained, and adjacent channel leak power is reduced.
FIG. 5 shows an explanation diagram when the distortion compensation processing is performed using the adaptive LMS in distortion compensator 9 shown in FIG. 4.
A symbol 15a is a multiplier for multiplying a transmission signal x(t) by a distortion compensation coefficient hn−1(p). This multiplier corresponds to the predistortion portion 91 shown in FIG. 4. Also, 15b is a transmission power amplifier having a distortion function f(p), and 15c is a feedback system in which feedback the output signal y(t) being output from transmission power amplifier 15b is performed. Also, 15d is a calculator (amplitude/power converter) for calculating a power p(=x2(t)) of the transmission signal x(t), and 15e is a distortion compensation coefficient storage (which corresponds to distortion compensation coefficient storage 90 shown in FIG. 4) for storing the distortion compensation coefficients each corresponding to each power of the transmission signal x(t).
Distortion compensation coefficient storage 15e outputs a distortion compensation coefficient hn−1(p) corresponding to the power p of the transmission signal x(t). Distortion compensation coefficient storage 15e also updates a distortion compensation coefficient hn−1(p) with distortion compensation coefficient hn(p) obtained by the LMS algorithm.
Further, 15f is a conjugate complex signal output portion, 15g is a subtractor outputting a difference e(t) between a transmission signal x(t) and a feedback demodulation signal y(t), 15h is a multiplier multiplying e(t) by u*(t), 15i is a multiplier multiplying hn−1(p) by y*(t), and 15j is a multiplier multiplying by a step size parameter μ, and 15k is an adder adding hn−1(p) to μe (t)u*(t). Also, 15m, 15n, 15p are delay portions by which a delay time D is added to the input signal. Here, the delay time D denotes a time duration from the time the transmission signal x(t) is input to the time the feed backed demodulation signal y(t) is input to subtractor 15g. 
Symbols 15f and 15h-15j constitute a rotation calculation section 16. A signal y(t) is the signal after being distorted. The delay time D being set in delay portions 15m, 15n, 15p is determined so as to satisfy D=D0+D1, where D0 is the delay time in transmission power amplifier 15b, and D1 is the delay time in feedback system 15c. 
When this delay time D is not set correctly, the distortion compensation function does not work effectively. Also, the greater the set error in the delay time is produced, the greater the leak power to the adjacent channels due to the sidelobe being produced occurs.
Using the above configuration, the following calculations are performed.hn(p)=hn−1(p)+μe(t)u*(t)e(t)=x(t)−y(t)y(t)=hn−1(p)x(t)f(p)u*(t)=x*(t)f*(p)=hn−1(p)y*(t)p=|x(t)|2 
Here, x, y, f, h, u, e are complex numbers, and * denotes a conjugate complex number.
Through the above calculation processing, the distortion compensation coefficient h(p) is updated so as to minimize the differential signal e(t) between the transmission signal x(t) and the feedbacked demodulation signal y(t). Finally, the value converges to an optimal distortion compensation coefficient, so that the distortion of the transmission power amplifier is compensated.
As described above, the principle of the distortion compensation apparatus is that feedback detection of a carrier wave obtained after quadrature modulation of the transmission signal is performed, the amplitudes of the transmission signal and the feedback signal are compared after digital conversion, and a distortion compensation coefficient is updated real time based on the above comparison result. According to this nonlinear distortion compensation system, it is possible to reduce distortion, and leak power as well, even through the operation is performed in a nonlinear region with high output, and also to improve the power load efficiency.
Now, in regard to the above setting of the delay time in the prior application, the applicant of the present invention has proposed one method, which is disclosed in the official gazette of the Japanese Unexamined Patent Publication No. 2001-189685. The method disclosed in the above patent publication is outlined below: A correlation value is calculated varying the phases between a transmission signal x(t) and a feedback signal. Based on the maximum value of this correlation, a total delay time produced in a distortion device (transmission power amplifier), a feedback loop, etc. is determined. The determined delay time is then set in each delay circuit for timing adjustment in the distortion compensation apparatus.
However, even once the delay time D is set correctly to satisfy D=D0+D1, in some cases, a stable and satisfactory distortion compensation operation may not be obtainable, and unnecessary outband radiation power may be produced.
This is caused by a clock jitter produced by thermal noise and other external disturbance in the analog system including an A/D converter and a D/A converter. Presence of the clock jitter causes an intense fluctuation in a feedback signal phase, and affects convergence of the distortion compensation coefficient.
The jitter produces repeated variations in the clock speed, to higher or lower. As a result, the feedback signal phases vary against the reference signal phases, as exemplarily shown in FIG. 6.
If such a phase variation caused by the clock jitter is not considered, the distortion compensation coefficient becomes unstably vibrated within the range of the phase variation. Because the distortion compensation coefficient is multiplied to the transmission signal, this causes unnecessary waves being produced.
Considering the above, in the prior application, which is disclosed in the PCT International Publication WO 03/103163, the applicant of the present invention has proposed the invention to enable a stable and satisfactory distortion compensation operation even when the phase difference between a reference signal and a feedback signal varies due to a jitter.
An exemplary embodiment of the invention disclosed in the prior application is shown in FIG. 7. In this FIG. 7, a distortion compensation coefficient lookup table (LUT) 61 is employed as a distortion compensation coefficient storage 15e (refer to FIG. 5), for storing distortion compensation coefficients corresponding to each power of transmission signals x(t).
Further, in FIG. 8, it is assumed that a phase difference φ is produced between the reference signal and the feedback signal, as shown by A, caused by the clock jitter. In such a case, if it is intended to correct this phase difference simply by detecting the phase difference φ between the reference signal and the feedback signal, phase correction cannot follow high-speed phase variation caused by the jitter.
Therefore, even when update of the distortion compensation coefficient lookup table 61 is performed through the phase correction, the distortion compensation coefficient cannot converge stably affected by a phase difference φpp, which impedes to obtain a satisfactory distortion compensation operation. Accordingly, in the invention disclosed in the prior application, an intermittent controller 69 is provided. With this, a phase correction period Δt and a distortion compensation coefficient update period ΔT are alternately generated.
The following method has been proposed in the prior invention: The phase difference φ between the reference signal and the feedback signal is corrected in the phase correction period Δt. Also, the distortion compensation coefficient is updated in the distortion compensation coefficient update period ΔT. The above operation is repeated thereafter.
More specifically, in the phase correction period Δt, the phase difference φ is measured for n times and averaged. Phase correction is then performed based on a mean phase difference. Further, in the distortion compensation coefficient update period ΔT having smaller phase difference than before as a result of the correction, the distortion compensation coefficient is updated at each clock.
Here, it is considered that the distortion compensation coefficient update period ΔT is sufficiently shorter than the period of phase variation.
As described above, according to the invention disclosed in the prior application, (i) the phase difference between the reference signal and the feedback signal is corrected; (ii) the distortion compensation coefficient is updated in the period when the phase difference becomes smaller as a result of the phase correction; (iii) update of the distortion compensation coefficient is suspended when the phase difference becomes greater, and instead, the phase difference is corrected; and (iv) thereafter, the distortion compensation coefficient is updated. Then, the above operation is repeated.
According to the invention in the prior application, the distortion compensation coefficient can be made to converge promptly without being affected by the phase difference, only by the effect of the phase difference of Δφ. Further, the distortion compensation coefficient update period is determined based on the phase difference between the reference signal and the feedback signal which is existent before the correction of the phase difference.
For example, the distortion compensation coefficient update period ΔT is set longer when the phase difference between the reference signal and the feedback signal is smaller, as shown by B. On the other hand, the distortion compensation coefficient update period ΔT is set shorter when the phase difference between the reference signal and the feedback signal is greater, as shown by C. With such a measure, when the phase difference is smaller, it becomes possible to make the distortion compensation coefficient converge promptly, because the update period can be set longer. In contrast, when the phase difference is greater, the distortion compensation coefficient update period becomes shorter, and the update of the distortion compensation coefficient is performed only in the period when the phase difference becomes smaller as a result of the correction.
Now, a further explanation will be given hereafter about the embodiment configuration (FIG. 7) according to the invention described in the patent document 2 mentioned earlier.
In FIG. 7, for a digital data group (transmission signals) forwarded from a transmission signal generator (not shown), distortion compensation processing is performed in distortion compensation apparatus 51, and is input to a D/A converter 52. This D/A converter 52 converts the digital transmission signal to an analog signal, and is input to a power amplifier 53 either directly or through a quadrature modulator and a frequency converter (which are not shown).
Power amplifier 53 amplifies the input signal and radiates to the air. The output of power amplifier 53 is input into an A/D converter 54 either directly or through a frequency converter and a quadrature demodulator (which are not shown). A/D converter 54 converts this input signal into a digital signal, and inputs the converted signal into a distortion compensation apparatus 51.
In distortion compensation apparatus 51, a distortion compensation coefficient lookup table (LUT) 61 stores a multiplicity of distortion compensation coefficients h(n) according to the power of each transmission signal x(t). A multiplier 62 multiplies each transmission signal by a distortion compensation coefficient h(n) corresponding to the transmission signal, and thus distortion compensation processing is performed.
An address generator 63 generates a readout address AR corresponding to the power of the transmission signal x(t). Address generator 63 then reads out a distortion compensation coefficient h(n) according to the above power, from distortion compensation coefficient lookup table 61, and inputs the readout distortion compensation coefficient h(n) into a multiplier 62.
Address generator 63 also generates a write address AW, and updates a distortion compensation coefficient by storing the distortion compensation coefficient h(n+1), which has been calculated in a distortion compensation coefficient updater 67, into distortion compensation coefficient lookup table 61. A delay circuit 64 outputs a reference signal x′(t) by delaying the input signal for a time duration from when the transmission signal x(t) is input to when a feedback signal y(t) is input to a subtractor 66. A complex multiplier 65 corrects the phase of the feedback signal y(t) so that the phase difference between the reference signal x′(t) and the feedback signal, which is output from A/D converter 54, becomes zero.
Subtractor 66 obtains a differential signal e(t) of between the reference signal x′(t) and the phase-corrected feedback signal y′(t). A distortion compensation coefficient updater 67 receives the differential signal e(t), and calculates a distortion compensation coefficient h(n+1) to reduce the above differential signal e(t), using an adaptive algorithm. Then, distortion compensation coefficient updater 67 updates the content h(n) of distortion compensation coefficient lookup table 61.
A phase adjustment circuit 68 detects a phase difference φ between the reference signal x′(t) and the feedback signal y′(t), and inputs the phase difference φ into complex multiplier 65. An intermittent controller 69 alternately generates a phase correction period Δt and a distortion compensation coefficient update period ΔT, and controls to perform a phase correction process and a distortion compensation coefficient update process alternately.
FIG. 9 shows a configuration diagram of a phase difference detector in phase adjustment circuit 68 shown in FIG. 7. Although not explicitly shown in FIG. 7, the transmission signal x(t) and the feedback signal y(t) are complex signals, and can be represented as follows:x(t)=Is+jQs y(t)=IF+jQF 
A quadrant detector 68a detects the quadrant in which a transmission signal x(t) is existent. A magnitude comparator 68b compares the magnitude of the real part with the imaginary part of the transmission signal x(t). Further, a vector existence angle range decider 68c decides in which section being divided on a 45-degree basis the transmission signal x(t) exists, based on the quadrant in which the transmission signal x(t) is existent and the comparison result of the magnitude, as shown in FIG. 10.
Similarly, a quadrant detector 68d detects the quadrant in which the feedback signal y(t) is existent. A magnitude comparator 68e compares the magnitude of the real part with the imaginary part. Further, a vector existence angle range decider 68f decides in which section being divided on a 45-degree basis the feedback signal y(t) exists, based on the quadrant in which the transmission signal x(t) is existent and the comparison result of the magnitude.
As such, a phase difference calculator 68g calculates the phase difference on a 45-degree basis, based on the sections of the transmission signal x(t) and the feedback signal y(t).
For example, assuming the transmission signal x(t) exists in a section IA, and the feedback signal y(t) exists in a section IIA, the phase difference is 90 degrees. An averaging section 68h calculates the mean value of the phase difference calculated in phase difference calculator 68g in the phase correction period, and sets this mean phase difference into complex multiplier 65.
As explained above, according to the invention described in the prior application (patent document 2), as shown in FIGS. 7 and 9, phase adjustment circuit 68 for exclusive use is needed.
However, according to the above-mentioned invention described in the prior application in the patent document 2, in order to obtain the phase difference, a multiplicity of circuits are to be constituted for the purposes of quadrant detection, magnitude comparison, and decision of the angle in which a vector exists. These circuits are not for general use and the cost becomes high.